A digital calibration algorithm with variable amplitude dithering for domain-extended pipeline ad cs

International Journal of VLSI design & Communication Systems (VLSICS) Vol.5, No.1, February 2014
DOI : 10.5121/vlsic.2014.5101 1
A DIGITAL CALIBRATION ALGORITHM WITH
VARIABLE-AMPLITUDE DITHERING FOR
DOMAIN-EXTENDED PIPELINE ADCS
Ting Li1
and Chao You2
1
Department of Electrical and Computer Engineering,
North Dakota State University, Fargo, USA
2
Information Engineering School, Nanchang University, Nanchang, China
ABSTRACT
The pseudorandom noise dither (PN dither) technique is used to measure domain-extended pipeline
analog-to-digital converter (ADC) gain errors and to calibrate them digitally, while the digital error
correction technique is used to correct the comparator offsets through the use of redundancy bits. However,
both these techniques suffer from three disadvantages: slow convergence speed, deduction of the amplitude
of the transmitting signal, and deduction of the redundancy space. A digital calibration algorithm with
variable-amplitude dithering for domain-extended pipeline ADCs is used in this research to overcome these
disadvantages. The proposed algorithm is implemented in a 12-bit, 100 MS/s sample-rate pipeline ADC.
The simulation results illustrate both static and dynamic performance improvement after calibration.
Moreover, the convergence speed is much faster.
KEYWORDS
Analog-to-digital Converter, Digital Calibration, Variable-amplitude Dithering
1. INTRODUCTION
The switched capacitor pipeline ADC is mainstream architecture in wireless communication and
digital consumer products. The performance of the switched capacitor pipeline ADC is
significantly limited by the linear errors due to comparator offsets, capacitor mismatches, and
finite operational-amplifier gain. A trend in modern ADC design is to utilize digital background
calibration to calibrate both capacitor mismatches and finite operational-amplifier gain [1] – [12].
In contrast, a 14-bit level performance without calibration has been achieved by carefully
matching the capacitors, using high open-loop gain operational-amplifiers and a multi-bit first
stage [13] [14]. Therefore, in order for a digital calibration to be useful, it should minimize the
analog circuits as well as the reasonable convergence time. Fortunately, digital circuits have
higher speeds and lower power consumption as compared to an analog counterpart [15].
Indeed, many digital calibration algorithms have been reported. For example, the parallel-ADC
algorithm uses a high accuracy but slow ADC to calibrate the errors of a high speed ADC [16].
This architecture requires an extra ADC, which costs both more power consumption and chip
area. On the other hand, the statistic-based algorithm [11] requires both a large memory to store
statistic data and an extremely low convergence speed. However, in the dither-based algorithm
[5, 6], the gain errors are modulated with a pseudorandom noise sequence (PN sequence) in the
analog domain and the gain errors are demodulated in the digital domain in order to extract the

2
errors from the processed input signal. In [12], the injection of dithers, together with the
comparator offsets, makes the output voltage of the current stage exceed the input range of the
following stage. This out-of-range input leads to a code loss. Two options are reported to prevent
an ADC from a code loss. The first option is to reduce the amplitude of the input signal, which
causes the transmitting signal’s signal-to-noise ratio (SNR) to decrease [12]. The other option is
to reduce the amplitude of the dither, which leads to a longer convergence time [17] [18].
To overcome these two conflicting issues, this paper introduces a variable-amplitude dither-based
algorithm to calibrate gain errors for domain-extended ADC. The proposed algorithm ensures
higher dither amplitude without reducing the amplitude of the input signal; therefore, the
convergence speed is much faster without harming the SNR of the transmitting signal. A similar
scheme is used in the traditional 1.5-bit/stage pipeline ADC to calibrate gain errors [9]. Moreover,
the algorithm is used in domain-extended pipeline ADCs and results in more space for the
comparator offsets.
This paper is organized as follows. The variable-amplitude dither-based digital background
calibration algorithm is described in Section 2. The proposed algorithm to correct all linear errors
is described in Section 3. Simulation results from the modelled 12-bit ADC are discussed in
Section 4 and conclusions are presented in Section 5.
2. VARIABLE-AMPLITUDE DITHER-BASED DIGITAL BACKGROUND
CALIBRATION ALGORITHM
Figure 1 shows the block diagram of a pipeline ADC using the traditional 1.5-bit/stage converter.
The pipeline ADC consists of multiple cascaded stages. Figure 2 shows the configuration of the
traditional 1.5-bit/stage ADC. Although the actual configuration is fully differential, the single-
ended configuration is shown for simplicity. In each stage, the converter works on two phases.
When Φଵ is high, it works on sample phase with the input simultaneously sampled on both
capacitors. In contrast, when Φଶ is high, it works on amplification phase with the input quantized
by the sub-ADC and the residue amplified to the full-scale of the next stage by a multiplying
digital-to-analog converter (MDAC) [19].
Figure 1. Pipeline ADC block diagram

3
Figure 2. Single stage converter in two phases
The performance of an ADC is highly dependent on the amplifier residue, because the comparator
offset can be corrected by the digital error correction technique using redundancy bits [20]. The
two major non-idealities which affect the amplifier residue are finite open-loop gains and
capacitor mismatches. The modelled feedback network is shown in Figure 3 with the real close-
loop gain G as [21]
G =
g
1 + gβ (1)
where g is the open-loop gain and β is the feedback factor.
Figure 3. Feedback network of a signal stage converter
When the open-loop gain approaches infinity,
G|୥→ஶ =
1
β
=
Cୱ
C୤
(2)
where Cୱ and C୤ are sample-and-hold capacitors as shown in Figure 1. According to Franco [21],
for a voltage-to-voltage converter,
ଵ
ஒ
equals the close-loop gain of the converter when the
amplifier is ideal.
In contrast to the ideal, the real close-loop gain G shown in Equation (1) can be factorized into

4
G =
g
1 + gβ
=
1
β
1
1 +
1
gβ
(3)
The real gain G shown in Equation (3) helps find the real amplifier residue of the current stages
V୰ୣୱ = G (V୧୬ −
1
2
Vୈ୅େ) (4)
Subscribing Equations (2) and (3) into Equation (4) yields
V୰ୣୱ =
1
β
1
1 +
1
gβ
൬V୧୬ −
1
2
Vୈ୅େ൰ =
Cୱ
C୤
1
1 +
C୤
gCୱ
(V୧୬ −
1
2
Vୈ୅େ)
(5)
Equation (5) takes both comparator mismatches and finite open-loop gain into consideration.
Therefore, both non-ideal factors that affect the amplifier residue can be combined into one error,
called the gain error. Thus, the real close-loop gain is
G୰ୣୟ୪ =
Cୱ
C୤
1
1 +
C୤
gCୱ
(6)
The equivalent model for a single stage ADC is shown in Figure 4.
Figure 4. Modelled single stage converter
Wang, Hurst, & Lewis [4] present the expression for the output word in the real condition as
D = D୰ +
1
G୰ୣୟ୪
D୆୉ (7)
where G୰ୣୟ୪ is the real close-loop gain, D୰ is the real output code, and D୆୉ is the output code of
the back-end ADC. The estimation of m =
ଵ
ୋ౨౛౗ౢ
is mୣ. Equation (7) can be rewritten as
D = D୰ + mୣD୆୉ (8)

5
D୆୉ = V୰ୣୱ = G୰ୣୟ୪ (V୧୬ − D୰) (9)
Subscribing Equation (9) into Equation (8) yields
D = D୰ + mୣ G୰ୣୟ୪ (V୧୬ − D୰) = V୧୬ (1 + e) − D୰e (10)
in which
e =
୫౛ି୫
୫
. (11)
e then, is the error coefficient in the estimate mୣ of m =
ଵ
ୋ౨౛౗ౢ
. In order to extract the error
coefficient e from Equation (11), one must use the following equation:
D୰ = Db + PN ΔA , (12)
where Dୠ is the output word of the sub-ADC without the injection of the PN sequence and ΔA is
the amplitude of the PN sequence. PN ∈{-1, 1} represents pseudorandom noise sequence.
Subscribing Equation (12) into Equation (10) yields
D = V୧୬ (1 + e) − Dbe − PN ΔAe. (13)
If the PN sequence uncorrelated with the input signal, then correlating D with PN can get the
error coefficient e. In order to fit the architecture used here, a new parameter is defined as
F(n) = D(n) − Db(n) (14)
where n is the number of conversion.
Because the Equations (10) and (12) include error coefficient e, the error coefficient e can then be
extracted from Equation (14). Subscribing both Equation (10) and Equation (12) into Equation
(14) yields
F(n) = (V୧୬(n) − Db(n))(1 + e) − PN(n)ΔAe. (15)
Correlating F with PN will give an estimate of error coefficient e; Therefore, multiplying
Equation (15) with PN yields
PN(n) · F(n) = PN(n)(V୧୬(n) − Db(n))(1 + e) − PNଶ
(n)ΔAe (16)
The next step is to calculate the average (represented by E[.]) with an infinite number of
cumulative sums. Since PN(n) ∈{-1, 1}, the first term in Equation (16) equals zero and yields

6
E[PN · F] = −ΔAe . (17)
As shown in Equation (17), the error coefficient e from the analog domain is extracted in the
digital domain. Adopting the least mean square (LMS) iterative calculation, mୣ keeps renewing
until it approaches m. The LMS iterative is shown as
mୣ(n) = mୣ(n − 1) + μPN(n − 1)F(n − 1) (18)
where μ is the step size for LMS iterative. For a smaller step size, longer convergence time
needed for an ADC to be stable. However, for a larger step size, larger stead-state errors existed
for an ADC.
3. CALIBRATION WITH VARIABLE-AMPLITUDE DITHERING FOR
DOMAIN-EXTENDED ARCHITECTURE
There are two constraints to the traditional technique. First, the dither amplitude constraint
originates from the trade-off between the dither amplitude and the signal amplitude. The signal
amplitude reduction leads to a decline of the SNR. Although a smaller dither amplitude makes the
signal amplitude larger, the smaller dither amplitude also requires longer convergence time.
Second, the redundancy space constraint results from the trade-off between the redundancy space
and the total amplitudes of the signal and the dither. The total amplitudes of the signal and the
dither reduction leads to either lower SNRs or longer convergence times. Although less
redundancy space makes the amplitude of the signal plus dither larger, it leaves less redundancy
to allow comparator offsets to be corrected.
A variable-amplitude dither used for domain-extended architecture is proposed to relieve both the
dither amplitude constraint and the redundancy space constraint. The amplitude of the dither is
variable according to the signal level. In this case, the amplitude of the dither is greatly increased
without a reduction of the amplitude of the signal; therefore, the convergence speed is much faster
without harming the SNR of the ADC. Moreover, the existence of more redundancy space in the
domain-extended architecture allows for the comparator offsets within a certain range to be
corrected.
The residue plot of a traditional 1.5-bit/stage ADC is shown in Figure 5 (a). The threshold
voltages are −
ଵ
ସ
V୰ୣ୤ and
ଵ
ସ
V୰ୣ୤. The input range is from −V୰ୣ୤ to V୰ୣ୤. Since the output of the
current stage is the input of the following stage of the pipeline ADC, the output range needs to be
the same as the input range to avoid a code loss. Therefore, like the input range, the output range
is also from −V୰ୣ୤ to V୰ୣ୤ . In contrast, the modified residue plot of a domain-extended 1.5-
bit/stage ADC is shown in Figure 5 (b). After two threshold voltages are added, the threshold
voltages are −
ଷ
ସ
V୰ୣ୤, −
ଵ
ସ
V୰ୣ୤,
ଵ
ସ
V୰ୣ୤, and
ଷ
ସ
V୰ୣ୤. The input range is from −
ହ
ସ
V୰ୣ୤ to
ହ
ସ
V୰ୣ୤. Like in
the traditional architecture, in order to prevent the ADC from a code loss in the modified
architecture, the output range needs to be the same as the input range: from −
ହ
ସ
V୰ୣ୤ to
ହ
ସ
V୰ୣ୤.

7
Figure 5. Residue plot of (a) a traditional 1.5-bit/stage ADC and (b) a domain-extended 1.5-bit/stage ADC
The domain-extended 1.5-bit/stage ADC allows for more space for comparator offsets, for higher
amplitude of the signal, for higher amplitude of the dither, as compared to the traditional 1.5-
bit/stage ADC. For domain-extended 1.5-bit/stage ADC, the total amplitudes of the signal and
the dither can be set within the −V୰ୣ୤ to V୰ୣ୤ range. Therefore, some space is available for
comparator offsets. However, in contrast, the traditional 1.5-bit/stage ADC needs the total
amplitudes of the signal and the dither to be set within the range of −0.8V୰ୣ୤ to 0.8V୰ୣ୤ in order to
have space for comparator offsets. The reduction of the total amplitudes of the signal and the
dither results in either a decline of SNR or a slower convergence speed.
The residue plot of the domain-extended 1.5-bit/stage ADC with variable-amplitude dithering is
shown in Figure 6 (a). The amplitude-variable dithers are injected between −
ଷ
ସ
V୰ୣ୤ to −
ଵ
ସ
V୰ୣ୤,
and three more comparators with threshold voltages equal to −
ହ
଼
V୰ୣ୤, −
ସ
଼
V୰ୣ୤, and−
ଷ
଼
V୰ୣ୤ are
added. A dither amplitude is chosen from −
ହ
ସ
V୰ୣ୤, −
ସ
ସ
V୰ୣ୤, −
ଷ
ସ
V୰ୣ୤, −
ଶ
ସ
V୰ୣ୤,
ଶ
ସ
V୰ୣ୤,
ଷ
ସ
V୰ୣ୤, and
ସ
ସ
V୰ୣ୤ depending on the PN value and the signal levels as shown in Figure 6(a). The total
amplitudes of the signal and the dither are within ±V୰ୣ୤, and so have room for comparator
redundancy. Therefore, the signal pulse dither between −
ଷ
ସ
V୰ୣ୤ and −
ଵ
ସ
V୰ୣ୤ is, in effect, a
constant-amplitude dither of ±
଻
଼
V୰ୣ୤ with a small signal within the range of ±
ଵ
଼
V୰ୣ୤, as shown in
Figure 6 (b). The actual signal stays unchanged, which means the SNR has not declined. The
dither amplitude is greatly increased, which leads to a fast convergence speed.

8
Figure 6. (a) Residue plot of the domain-extended 1.5-bit/stage ADC with variable-amplitude dithering, (b)
equivalent constant-amplitude PN dithering of (a), and (c) residue plot of the domain-extended 1.5-bit/stage
ADC with constant-amplitude dithering
However, for the domain-extended 1.5-bit/stage ADC with constant-amplitude dithering, the
dither amplitude is only
ଵ
ଶ
V୰ୣ୤ as shown in Figure 6 (c). As mentioned above, in the case of one
signal level divided into four sub-levels, the dither amplitude is increased from
ଵ
ଶ
V୰ୣ୤ to
଻
଼
V୰ୣ୤
without reducing the signal amplitude. Therefore, the variable-amplitude dithering calibration
allows higher amplitude of the dither without the decline of SNR.
In dither-based architecture shown in previous studies [9] [15], when the signal stays at a high
level all the time, more dither cannot be added. Therefore, the error coefficient e cannot be
extracted in the digital domain according to Equation (16), (17), and (18). This is because the first
term of Equation (16) equals zero after an infinite number of cumulative sums and the average is
calculated. The second term always equals zero because ΔA equals zero. Therefore, the LMS
iterative is not applicable for this condition. This circumstance constrains the application.
However, the proposed variable-amplitude dithering does not suffer from the limitation. The
proposed variable-amplitude dithering also offers substantial savings in convergence times, even
though the signal stays at high level at all times. This works because the signal stays at its highest
level, the PN=-1, and the amplitude of the dither is
ଵ
ଶ
V୰ୣ୤, which is the smallest dither amplitude
in the proposed method.

9
These concepts are realized on the circuit level in the following concrete example. A MDAC
works in two phases: the sample phase and the amplification phase. During the sample phase, the
sample input signal is administered to all capacitors. Figure 7 shows how the MDAC works in the
amplification phase with a signal range from −
ଷ
ସ
V୰ୣ୤ to −
ଵ
ସ
V୰ୣ୤. In this configuration, V୰ୣ୤ equals
half of the power supply voltage Vୢୢ. Variable-amplitude dithering is realized by adding three
more comparators and splitting a capacitor into four capacitors, cଵ, cଶ, cଷ, and cସ with each
capacitor valued at
ୡ౜
ସ
. The amplitudes of the dithers, in all sub-levels of the main level from
−
ଷ
ସ
V୰ୣ୤ to −
ଵ
ସ
V୰ୣ୤ , are shown in Table 1. Dither injections are controlled through switches
dependant on both the output of the encoder and the PN value. The following examples lay out
the various permutations for the dither injections of the main level. To inject dither of
ହ
ସ
V୰ୣ୤, cଵ is
switched to −Vୢୢ , and the other three capacitors are switched to −V୰ୣ୤ for the signal range
between −
ଷ
ସ
V୰ୣ୤ ~ −
ହ
଼
V୰ୣ୤, if PN is -1. To inject dither of −
ଵ
ଶ
V୰ୣ୤, both cଵ and cଶ are switched
to V୰ୣ୤, and the other two capacitors are switched to 0 for the signal range between −
ଷ
ସ
V୰ୣ୤ ~ −
ହ
଼
V୰ୣ୤, if PN is 1. For the other main level: −
ହ
ସ
V୰ୣ୤ ~ −
ଷ
ସ
V୰ୣ୤ and −
ଵ
ସ
V୰ୣ୤ ~
ହ
ସ
V୰ୣ୤, the dither
injection method is similar as the main level: −
ଷ
ସ
V୰ୣ୤ ~ −
ଵ
ସ
V୰ୣ୤.
Figure 7. MDAC architecture in amplification phase with the signal range from −
ଷ
ସ
V୰ୣ୤ to −
ଵ
ସ
V୰ୣ୤
Table 1 Amplitude of the dithers in all sub-levels of the main level from −
ଷ
ସ
V୰ୣ୤ to −
ଵ
ସ
V୰ୣ୤
‫܄‬ܑ‫ܖ‬ Amplitude of dithers
Main level Sub-levels PN=-1 PN=1
−
ଷ
ସ
V୰ୣ୤ ~ −
ଵ
ସ
V୰ୣ୤ −
ଷ
ସ
V୰ୣ୤ ~ −
ହ
଼
V୰ୣ୤
5
4
V୰ୣ୤ −
1
2
V୰ୣ୤
−
ହ
଼
V୰ୣ୤ ~ −
ଵ
ଶ
V୰ୣ୤
V୰ୣ୤
−
3
4
V୰ୣ୤
−
ଵ
ଶ
V୰ୣ୤ ~ −
ଷ
଼
V୰ୣ୤
3
4
V୰ୣ୤
−V୰ୣ୤
−
ଷ
଼
V୰ୣ୤ ~ −
ଵ
ସ
V୰ୣ୤
1
2
V୰ୣ୤ −
5
4
V୰ୣ୤

10
4. SIMULATION RESULTS
To illustrate the effectiveness of the digital calibration with the variable-amplitude dithering for
the domain-extended architecture, a 12-bit pipeline ADC is simulated in Matlab. This ADC
consists of ten 1.5-bit/stage converters followed by a 2-bit flash ADC with digital error
correction, digital calibration, and PN sequence generation as shown in Figure 8. The gain errors
and comparator offsets in all stages are chosen between 2% to 5%. The initial calibration
coefficient mୣ set to be its standard value, which is
ଵ
ୋ
= 0.5. Full-scale sinusoid is used as the test
input. The linear gain calibration is applied to the first four stages, and the comparator offset
corrections are applied to all 1.5-bit/stage converters.
Figure 8. Block diagram of the system architecture
The convergence of the error coefficient e =
୫౛ି୫
୫
for a 12-bit pipeline ADC using the proposed
calibration with variable-amplitude dithering is shown in Figure 9. Convergence is reached after
approximately 8 × 10ହ
samples in this case. However, convergence is reached after
approximately 4 × 10଺
samples in the calibration with constant-amplitude dithering [12]. At a
conversion rate of 75 MS/s, the ADC using the proposed calibration with variable-amplitude
dithering needs a time constant of approximately 2.2 ms, while the ADC using the calibration
with constant-amplitude dithering needs a time constant of approximately 11 ms. With the
increasing of ADC resolution, the LMS iterative step size μ needs to be reduced; therefore, the
calibration time constant is increased accordingly. Appling the proposed calibration with variable-
amplitude dithering to the high resolution ADCs can greatly reduce the convergence time.
Figure 9. Convergence of the error coefficient

11
A full-scale sinusoidal input with low frequency is used to test the static performance. Figure 10
(a) and Figure 10 (b) show the differential nonlinearity (DNL), of a 12-bit pipeline ADC, without
calibration and with calibration, respectively. The peak DNL is reduced from 3.1 LSB in the un-
calibrated case to 0.5 LSB in the calibrated case. Figure 11 (a) and Figure 11 (b) show the
integration nonlinearity without calibration and with calibration, respectively. The peak INL is
reduced from 31 LSB in the un-calibrated case to 0.5 LSB in the calibrated case.
Figure 10. 12-bit ADC DNL: (a) without calibration and (b) with calibration
Figure 11. 12-bit ADC INL: (a) without calibration and (b) with calibration
A full-scale sinusoidal with an input frequency of 45 MHz and a sample-rate of 100 MS/s are
used to test the dynamic performance. The output spectrum of the ADC without calibration is
shown in Figure 12 (a). The spurious-free dynamic range (SFDR) in this un-calibrated case is 51
dB, while the SNR is only 40.4 dB, which corresponds to an effective number of bits (ENOB) of
6.4. Also, in this case, the total harmonic ratio (THD) is 51 dB. The output spectrum of the ADC
with the proposed calibration is shown in Figure 12 (b). The tone is reduced, and the SFDR is
87.7 dB. Also the noise floor is lowered, which shows a 69.3 dB SNR with an ENOB of 11.2. The
THD is 86.2 dB in the calibrated case. Table 2 shows the performance of the 12- bit ADC without
calibration and with calibration.
Figure 12. 12-bit ADC output spectrum: (a) without calibration and (b) with calibration
(a) (b)
(a) (b)

12
Table 2. Performance of a 12-bit ADC
Parameter Without
calibration
With
calibration
Sample-
rate(MS/s)
100 100
Resolution(bit) 12 12
DNL (LSB) -1~3.1 -0.4~0.5
INL (LSB) -31~30.8 -0.5~0.4
SFDR(dB) 51 87.7
SNR(dB) 40.4 69.3
THD(dB) 51 86.2
ENOB(bit) 6.4 11.2
5. CONCLUSION
A digital calibration with variable-amplitude dithering is applied to a domain-extended pipeline
ADC. This technique allows for large average amplitude dither injections without scarifying the
signal amplitude, which means fast convergence speed while keeping the SNR not changed. For
the dither-based calibrations [9] [15], the calibrations cannot proceed if the signal stays at high
level all the time. This constrain limits the application. The introduction of domain-extended
architecture allows the digital calibration to proceed even though the signal stays at high level all
the time. Also, the redundancy space plus the total amplitude of the signal and the dither is
limited by the quantify range. The domain-extended architecture extends the quantify range,
which results in higher SNR, faster convergence speed, or more redundancy space for the
comparator offsets to be corrected. The simulated ADC demonstrates good dynamic and static
performance within a practical time without paying significant penalty in the circuit complexity.
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AUTHORS
Ting Li received the B.S. degree and the M.S. degree from China University of Mining
of Technology, Beijing, China, in 2008 and 2010 respectively. Since August 2010, she
has been with North Dakota State University (NDSU), Fargo, North Dakota, USA, where
she is working toward the Ph.D. degree in electrical engineering.
present, she worked with NDSU as
on accuracy enhancement techniques in pipeline analog
current research interest include analog/mixed
analog/mixed-signal integrated-circuit design.
Chao You received the B.S. degree from Nankai University in 1999. He received the
M.S. degree and the Ph.D. degree from Rensselaer Polytechnic Institute, Troy, New
York, USA, in 2003 and 2005, respectively. From August 2005 to 2011, he worked
with NDSU as an Assistant Professor. From 2011 to present, he worked with NDSU as
an Associate Professor. He is also a Visiting Professor in Nanchang University,
Nanchang, Jiangxi, China. His research is very large scale integration.
International Journal of VLSI design & Communication Systems (VLSICS) Vol.5, No.1, Februa
W. Yang, D. Kelly, I. Mehr, M. T. Sayuk, and L. Singer, “A 3-V 340-mW 14-b 75-Msample/s CMOS
dB SFDR at Nyquist input,” IEEE Journal of Solid-State Circuits, vol. 36, no. 12, pp.
A. M. A. Ali, C. Dillon, R. Sneed, A. S. Morgan, S. Bardsley, J. Kornblum, and L. Wu, “A 14
MS/s IF/RF sampling pipelined ADC with 100 dB SFDR and 50 fs jitter,” IEEE Journ
State Circuits, vol. 41, no. 8, pp. 1848–1855, Aug. 2006.
S. Yang, J. Cheng, and P. Wang, “Variable-amplitude dither-based digital background calibration
order nonlinear error in pipelined ADCs,” Microelectronics Journal, vol.
C. Tsang, Y. Chiu, J. Vanderhaegen, S. Hoyos, C. Chen, R. Brodersen, and B. Nikolic, “Background
304, 2008.
A. Panigada, I. Galton. “Digital background correction of harmonic distortion in pipelined ADCs,”
IEEE Trans. on Circuits and Systems I: Regular Papers, vol. 53, pp. 1885–1895, 2006.
A. Meruva, and B. Jalali, “Digital background calibration of higher order nonlinearities in pipelined
ADCs,” IEEE International Symposium on Circuits and Systems, pp. 1233–1236, 2007.
H. Lee, and M. F. Tompsett, “A 10-b 15-MHz CMOS recycling two
of Solid-State Circuits, vol. 25, no. 6, pp. 1328–1338, Dec. 1990.
S. H. Lewis, P. R. Gray, “A pipelined 5-Msample/s 9-bit analog-to-digital converter,” IEEE Journal
State Circuits, vol. 22, no. 6, pp. 954–961, 1987.
tional amplifier fundamentals. In Design with Operational Amplifiers and Analog
Integrated Circuits,” 3rd ed., Publisher: McGraw-Hill Science, USA, pp. 1-60, 2001.
degree and the M.S. degree from China University of Mining
of Technology, Beijing, China, in 2008 and 2010 respectively. Since August 2010, she
has been with North Dakota State University (NDSU), Fargo, North Dakota, USA, where
.D. degree in electrical engineering. From August 2010 to
present, she worked with NDSU as a Teaching Assistant. Her doctoral research focuses
on accuracy enhancement techniques in pipeline analog-to-digital converter design. Her
include analog/mixed-signal behavioral simulation and
circuit design.
received the B.S. degree from Nankai University in 1999. He received the
M.S. degree and the Ph.D. degree from Rensselaer Polytechnic Institute, Troy, New
York, USA, in 2003 and 2005, respectively. From August 2005 to 2011, he worked
tant Professor. From 2011 to present, he worked with NDSU as
an Associate Professor. He is also a Visiting Professor in Nanchang University,
Nanchang, Jiangxi, China. His research is very large scale integration.
ICS) Vol.5, No.1, February 2014
13
Msample/s CMOS
State Circuits, vol. 36, no. 12, pp.
A. M. A. Ali, C. Dillon, R. Sneed, A. S. Morgan, S. Bardsley, J. Kornblum, and L. Wu, “A 14-bit 125
MS/s IF/RF sampling pipelined ADC with 100 dB SFDR and 50 fs jitter,” IEEE Journal of Solid-
based digital background calibration
onics Journal, vol.
C. Tsang, Y. Chiu, J. Vanderhaegen, S. Hoyos, C. Chen, R. Brodersen, and B. Nikolic, “Background
A. Panigada, I. Galton. “Digital background correction of harmonic distortion in pipelined ADCs,”
nd calibration of higher order nonlinearities in pipelined
1236, 2007.
MHz CMOS recycling two-step A/D
1338, Dec. 1990.
digital converter,” IEEE Journal
tional amplifier fundamentals. In Design with Operational Amplifiers and Analog

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